Apparatus for estimating and correcting baseband frequency error in a receiver

ABSTRACT

An apparatus for estimating and correcting baseband frequency error in a receiver is disclosed. An equalizer performs equalization on a sample data stream and generates filter tap values based on the equalization. An estimated frequency error signal is generated based on at least one of the filter tap values. A rotating phasor is generated based on the estimated frequency error signal. The rotating phasor signal is multiplied with the sample data stream to correct the frequency of the sample data stream. Alternatively, a channel estimator performs channel estimation and generates Rake receive finger weights based on at least one of the finger weights. An estimated frequency error signal is generated based on at least one of the finger weights.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.12/688,242, filed Jan. 1, 2010, which is a continuation-in-part of U.S.patent application Ser. No. 12/265,929, filed Nov. 6, 2008, which is acontinuation of U.S. patent application Ser. No. 11/209,097, filed Aug.22, 2005, now U.S. Pat. No. 7,457,347, issued Nov. 25, 2008, whichclaims the benefit of U.S. Provisional Patent Application No.60/625,874, filed Nov. 8, 2004; and is a continuation-in-part of U.S.patent application Ser. No. 12/512,203, filed Jul. 30, 2009, which is acontinuation of U.S. patent application Ser. No. 11/265,373, filed Nov.2, 2005, now U.S. Pat. No. 7,570,690, issued Aug. 4, 2009, which claimsthe benefit of U.S. Provisional Patent Application No. 60/625,188, filedNov. 5, 2004, which are incorporated by reference as if fully set forth.

FIELD OF INVENTION

The present invention is related to wireless receivers. Moreparticularly, the present invention is related to a method and apparatusfor estimating and correcting frequency error at baseband in a receiver.

BACKGROUND

Adaptive receivers, such as a normalized least mean square (NLMS)equalizer used in wireless transmit/receive units (WTRUs) and basestations, optimize their associated filter tap values through aniterative procedure that requires multiple iterations to nearconvergence. The tap values converge as time passes to a minimum meansquare error (MMSE) solution used to perform channel estimation.

An NLMS receiver includes an equalizer having an equalizer filter whichis continually in the process of converging as it tries to track atime-varying channel. The more complex it is to track the channel, thefurther the tap values of the equalizer will be from convergence.Generally, faster channels (i.e., channel states that evolve rapidly)are difficult for the equalizer to track. Residual automatic frequencycontrol (AFC) errors in the baseband input into the equalizer causechannels to appear faster than they really are. The increase in theapparent speed of the channel can only be partially mitigated byincreasing the step-size of an NLMS algorithm implemented by the NLMSreceiver. The increased step-size allows the equalizer filter to moreaccurately track “fast” channels, but it also increases errors in theMMSE solution which cause degradation in the performance of thereceiver.

Receivers that employ channel estimation are also degraded by residualAFC errors. Since the bandwidth of the appropriate equalizer filter usedin channel estimation is a function of the apparent speed of thechannel, large AFC errors force the use of wide-band filters that do notefficiently suppress noise, thus leading to less accurate channelestimates. A simple solution is desired to suppress the residual AFCerrors.

SUMMARY

The present invention is related to an apparatus for estimating andcorrecting baseband frequency error in a receiver. In one embodiment, anequalizer performs equalization on a sample data stream and generatesfilter tap values based on the equalization. An estimated frequencyerror signal is generated based on at least one of the filter tapvalues. A rotating phasor is generated based on the estimated frequencyerror signal. The rotating phasor signal is multiplied with the sampledata stream to correct the frequency of the sample data stream. Inanother embodiment, a channel estimator performs channel estimation andgenerates Rake receiver finger weights based on at least one of thefinger weights. An estimated frequency error signal is generated basedon at least one of the finger weights.

BRIEF DESCRIPTION OF THE DRAWINGS

A more detailed understanding of the invention may be had from thefollowing description, given by way of example and to be understood inconjunction with the accompanying drawings wherein:

FIG. 1 is a block diagram of an example BFC system including a frequencyerror estimator for removing residual AFC errors in accordance with oneembodiment of the present invention;

FIG. 2 is a block diagram of the frequency error estimator of the system100 of FIG. 1;

FIG. 3 is a block diagram of an example BFC system in accordance withanother embodiment of the present invention;

FIG. 4 is a high level flow diagram of a process for correcting thefrequency of a sample data stream in a wireless communication receiverhaving an equalizer that performs equalization in accordance with oneembodiment of the present invention;

FIG. 5 is a flow diagram of a process for generating the estimatedfrequency error signal based on a filter tap value extracted from filtertap values generated by the equalizer used in the process of FIG. 4;

FIG. 6 is a flow diagram of a process for generating the estimatedfrequency error signal based on a plurality of extracted tap values thatare averaged in accordance with one embodiment of the present invention;

FIG. 7 is a flow diagram of a process for comparing the magnitude of thephase difference signal with the value of the threshold signal todetermine whether the estimated frequency error signal should beprevented from being updated in accordance with one embodiment of thepresent invention;

FIG. 8 is a flow diagram of a process for comparing the instantaneouspower of the phase difference signal with the value of the thresholdsignal to determine whether the estimated frequency error signal shouldbe prevented from being updated in accordance with one embodiment of thepresent invention;

FIG. 9 is a high level flow diagram of a process for correcting thefrequency of a sample data stream in a wireless communication receiverhaving a channel estimator that performs channel estimation inaccordance with one embodiment of the present invention; and

FIG. 10 is a flow diagram of a process for generating the estimatedfrequency error signal based on a finger weight extracted from Rakereceiver finger weights generated by the channel estimator used in theprocess of FIG. 9.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The preferred embodiments will be described with reference to thedrawing figures where like numerals represent like elements throughout.

Hereafter, the terminology “WTRU” includes but is not limited to a userequipment (UE), a mobile station, a laptop, a personal data assistant(PDA), a fixed or mobile subscriber unit, a pager, or any other type ofdevice capable of operating in a wireless environment. When referred tohereafter, the terminology “base station” includes but is not limited toan access point (AP), a Node-B, a site controller or any other type ofinterfacing device in a wireless environment.

The features of the present invention may be incorporated into anintegrated circuit (IC) or be configured in a circuit comprising amultitude of interconnecting components.

Hereinafter, the present invention will be described in terms of theNLMS equalizer. However, it should be noted that the NLMS equalizerbased receiver is provided as an example and the present invention canbe applied to receivers implementing any other adaptive equalizationalgorithm and to receivers employing channel estimation such as blockbased equalizers and rake receivers.

FIG. 1 is a block diagram of an exemplary BFC system 100 for removingresidual AFC errors in accordance with one embodiment of the presentinvention. The BFC system 100 may be incorporated in a WTRU or a basestation. The BFC system 100 includes a multiplier 102, an equalizer 104,a frequency error estimator 106, a controller 108 and a numericallycontrolled oscillator (NCO) 110. The equalizer 104 processes a sampledata stream 112 provided by a receiver front end (not shown) via themultiplier 102. The equalizer may operate in accordance with an NLMSalgorithm. However, any other type of adaptive equalizer algorithm maybe applied.

Filter tap values 114 generated by the equalizer 104 are provided as aninput to the frequency error estimator 106. The frequency errorestimator 106 generates an estimated frequency error signal 116. Theresidual frequency errors after AFC can be greatly reduced by BFC basedsolely on observation of at least one tap value in the equalizer 104, oralternatively from partial channel estimates, such as a Rake fingercomplex weight estimation. BFC is accomplished by estimating thefrequency error based on observation of the one or more taps in theequalizer 104, generating a correction signal consisting of a complexsinusoid (or rotating phasor), correcting the input samples data streamby multiplying it by the phasor and applying frequency corrected samples118 to the input of the equalizer 104 in a closed loop fashion.

The residual frequency error is estimated by periodically measuring thephase change of one or more of the tap values of the equalizer 104 (oralternatively, partial channel estimates). Much of the phase changemeasured on the equalizer filter taps 114 from sample to sample is dueto noise and fading. However, phase changes due to fading and noise arezero mean (e.g., have a mean value of zero). Therefore, the expectedvalue of any sample average will be zero, i.e., the average value of thesignal is zero. Thus, filtering can be used to remove noise and fadingcomponents which cause phase change from the overall phase changes (dueto, e.g., residual AFC errors) and to recover the slowly varying phasechange due to the frequency error.

Once the frequency error is estimated by the frequency error estimator106, the controller 108 processes the estimated frequency error signal116 to generate a frequency adjustment signal 120. The controller 108may simply adjust the gain of the estimated frequency error signal 116or may process the estimated frequency error signal 116 with a morecomplicated algorithm (e.g., a proportional-integral-derivative (PID)).The frequency adjustment signal 120 is fed to the NCO 110 whichgenerates a rotating phasor 122 which corresponds to the frequencyadjustment signal 120. The multiplier 102 multiplies the rotating phasor122 with the sample data stream 112 to generate the frequency correctedsamples 118 input into the equalizer 104.

Residual AFC errors manifest themselves in the baseband as amultiplicative error in the baseband signal and has the form of acomplex sinusoid, such as g(t)*exp(j*2pi*f*t) where g(t) is the desireduncorrupted baseband signal and exp(j*2pi*f*t) is the complex sinusoidrepresenting the error. By multiplying by exp(−j*2pi*f*t), the complexsinusoids cancel leaving only the desired signal g(t). The estimatedfrequency error signal 116 is input to the controller 108 which, inturn, outputs a signal 120 which may be, for example, a scaled (i.e.,proportional) version of the input, e.g., four times the value of theestimated frequency error signal 116. The output signal 120 of thecontroller 108 may also include other terms such as a term proportionalto the integrals and/or derivatives of the estimated frequency errorsignal 116. More generally, the output signal 120 could also be clippedto be within some range or have other such non-linear function appliedto it. The NCO (110) takes as an input a frequency value and outputs aconstant magnitude complex signal with instantaneous frequency equal tothe value of the input, e.g., exp(j*2pi*f*t), where f is the inputfrequency.

FIG. 2 is a block diagram of the frequency error estimator 106 of theBFC system 100 shown in FIG. 1. The frequency error estimator 106includes a tap extraction unit 202, a delay unit 204, a conjugategenerator 206, multipliers 208, 210, an arctangent unit 212, a magnitudedetector 214, an averaging filter 216, a phase change filter 218 and acomparator 220. The equalizer generates filter taps 114 which aresupplied to the frequency error estimator 106.

In the frequency error estimator 106, the tap extraction unit 202extracts and outputs an appropriate tap value or average of tap valuesonto an output signal 203 from the filter taps 114, (or alternativelyfrom a channel estimator), to use for performing frequency estimation.For example, at least one appropriate tap value corresponding to an FSPin a particular channel may be extracted from the equalizer filter taps114. The tap extraction unit 202 may also track frequency drifting ofthe extracted tap value.

The extracted tap value 203 is forwarded to a delay unit 204 and aconjugate unit 206. The delay unit 204 delays the extracted tap value203 for a predetermined period of time by outputting a delayed tap value205. The conjugate generator is used to generate a conjugate 207 of theextracted tap value 203. The multiplier 208 multiplies the delayed tapvalue 205 by the conjugate tap value 207. The output 209 of themultiplier 208 has a phase value equal to the phase difference betweenthe delayed tap value 205 and the conjugate tap value 207. This phasevalue is proportional to the average frequency of the signal 203 andtherefore of the sample data stream 112.

The arctangent unit 212 measures an angle value 213 of the output 209 ofthe multiplier 208. The angle value 213 is equal to the phase differencebetween signal 205 and signal 207. Averaging the angle value 213 istherefore equivalent to averaging the phase difference between signal205 and signal 207. The angle value 213 is filtered by the phase changefilter 218 for averaging the angle value 213. The measured average phasedifference and the known delay are used to generate the estimatedfrequency error signal 116.

For example, with a delay D (sec) and phase measured in radians, thegain of the frequency error estimator 106 is 1/(2*PI*D). The “gain”refers to the conversion of a signal with a net frequency error, (asindicated by signal 114), to an observed value of the estimatedfrequency error signal 116. If the signal 114 has an average frequencyof 1 Hz, then the output value on the estimated frequency error signal116 will be 1/(2*PI*D).

The magnitude detection unit 214 calculates the magnitude of the output209 of the multiplier 208 and sends a calculated magnitude value 215 toa first input, X, of the comparator 220 and to the averaging filter 216for averaging. The multiplier 210 multiplies the output signal 217 ofthe averaging filter 216 (i.e., the average value of signal 215) with athreshold factor value 219 (e.g., a scaling factor having a value T) togenerate a threshold signal 222 which is sent to a second input, Y, ofthe comparator 220. The value of the threshold signal 222 may be set toa fraction of the average amplitude of the output 209 of the multiplier208. The threshold factor value, T, may be set, for example, to ⅓. Thecomparator 220 compares the calculated magnitude value 215 with thevalue of the threshold signal 222 and sends a hold signal 221 to thephase change filter 218 if the calculated magnitude value 215 is belowthe value of the threshold signal 222.

The magnitude of the output 209 of the multiplier 208 may be measuredand compared to the average amplitude of the output 209 of themultiplier 208, whereby the phase change filter 218 is paused wheneverthe magnitude of the output 209 of the multiplier 208 drops below athreshold. When the filter 218 is paused, the estimated frequency errorsignal 116 does not change (i.e., the signal 116 is not updated), theinput 213 is not used, and the internal state of the filter 218 does notchange. The hold signal 221 is true whenever the signal 209 isrelatively small. This has the effect of discarding the angle values onsignal 213 whenever they are noisiest, and improving the estimatedfrequency error signal 116 when the channel undergoes deep fades.

Alternately, a power detector (not shown) may be substituted for themagnitude detector 214 to calculate the average power (i.e., the squaredmagnitude) of the output 209 of the multiplier 208, whereby theinstantaneous power of the output 209 is compared to some fraction ofthe average power. Other variations are also possible.

FIG. 3 is a block diagram of a BFC system 300 for removing residual AFCerrors in accordance with another embodiment of the present invention.The BFC system 100 may be incorporated in a WTRU or a base station. TheBFC system 300 includes a multiplier 302, a rake combiner (or a blockequalizer) 304, a channel estimator 306, a frequency error estimator308, a controller 310, and an NCO 312.

In a Rake receiver, a finger weight is determined based on the channelestimation on a particular multipath component assigned to a Rakefinger. The channel estimator 306 generates Rake receiver finger weights316 which are provided as an input to the frequency error estimator 308.The frequency error estimator 308 operates in a fashion similar to thefrequency error estimator 106 shown in FIGS. 1 and 2.

Once the frequency error is estimated by the frequency error estimator308, the controller 310 processes the estimated frequency error signal318 to generate a frequency adjustment signal 322. The frequencyadjustment signal 322 is fed to the NCO 312 which generates a rotatingphasor 324 which corresponds to the frequency adjustment signal 322. Themultiplier 302 multiplies the rotating phasor 324 with the sample datastream 314 to generate the frequency corrected samples 320 input intothe channel estimator 306 and the Rake combiner 304. Alternatively, ablock equalizer may be used instead of the Rake combiner 304.

The estimated frequency error signal 318 generated by the frequencyerror estimator 308 is processed by the controller 310 and the NCO 312which applies a rotating phasor 324 to the sample data stream 314.

FIG. 4 is a high level flow diagram of a process 400 including methodsteps for correcting the frequency of a sample data stream in a wirelesscommunication receiver having an equalizer that performs equalization inaccordance with one embodiment of the present invention. In step 405,equalization is performed on a sample data stream. In step 410, filtertap values are generated based on the equalization. In step 415, anestimated frequency error signal is generated based on at least one ofthe filter tap values. In step 420, a rotating phasor signal isgenerated based on the estimated frequency error signal. In step 425,the rotating phasor signal is multiplied with the sample data stream tocorrect the frequency of the sample data stream.

FIG. 5 is a flow diagram of a process 500 including method steps forgenerating the estimated frequency error signal based on a filter tapvalue extracted from filter tap values generated by the equalizer usedin process 400 of FIG. 4. In step 505, an appropriate tap value isextracted from the filter tap values (generated in step 410 of FIG. 4).In step 510, the extracted tap value is delayed. In step 515, aconjugate of the extracted tap value is generated. In step 520, theconjugate of the extracted tap value is multiplied with the delayedextracted tap value to generate a phase difference signal whichrepresents the phase difference between the conjugate extracted tapvalue and the delayed extracted tap value. In step 525, the phasedifference between the conjugate extracted tap value and the delayedextracted tap value is measured. In step 530, the estimated frequencyerror signal is generated by averaging the measured phase difference. Instep 535, the estimated frequency error signal is selectively preventedfrom being updated based on a value of a threshold signal.

FIG. 6 is a flow diagram of a process 600 including method steps forgenerating the estimated frequency error signal based on a plurality ofextracted tap values that are averaged in accordance with one embodimentof the present invention. In step 605, a plurality of tap values areextracted from the filter tap values (generated in step 410 of FIG. 4).In step 610, the extracted tap values are averaged to generate anaverage value of the extracted tap values. In step 615, the average tapvalue is delayed. In step 620, a conjugate of the average tap value isgenerated. In step 625, the conjugate of the average tap value ismultiplied with the delayed average tap value to generate a phasedifference signal which represents the phase difference between theconjugate average tap value and the delayed average tap value. In step630, the phase difference between the conjugate average tap value andthe delayed average tap value is measured. In step 635, the estimatedfrequency error signal is generated by averaging the measured phasedifference. In step 640, the estimated frequency error signal isselectively prevented from being updated based on a value of a thresholdsignal.

FIG. 7 is a flow diagram of a process 700 including method steps forcomparing the magnitude of the phase difference signal with the value ofthe threshold signal to determine whether the estimated frequency errorsignal should be prevented from being updated in accordance with oneembodiment of the present invention. In step 705, a magnitude of thephase difference signal is calculated. In step 710, the magnitude of thephase difference signal is averaged. In step 715, the threshold signalis generated by multiplying a scaling factor with the averaged magnitudeof the phase difference signal. In step 720, the magnitude of the phasedifference signal is compared with the value of the threshold signal. Instep 725, the estimated frequency error signal is prevented from beingupdated if the magnitude of the phase difference signal is below thevalue of the threshold signal.

FIG. 8 is a flow diagram of a process 800 including method steps forcomparing the instantaneous power of the phase difference signal withthe value of the threshold signal to determine whether the estimatedfrequency error signal should be prevented from being updated inaccordance with one embodiment of the present invention. In step 805,the instantaneous power of the phase difference signal is calculated. Instep 810, the instantaneous power of the phase difference signal isaveraged. In step 815, the threshold signal is generated by multiplyinga scaling factor with the averaged instantaneous power of the phasedifference signal. In step 820, the instantaneous power of the phasedifference signal is compared with the value of the threshold signal. Instep 825, the estimated frequency error signal is prevented from beingupdated if the instantaneous power of the phase difference signal isbelow the value of the threshold signal.

FIG. 9 is a high level flow diagram of a process 900 including methodsteps for correcting the frequency of a sample data stream in a wirelesscommunication receiver having a channel estimator that performs channelestimation in accordance with one embodiment of the present invention.In step 905, channel estimation is performed on a sample data stream. Instep 910, Rake receiver finger weights are generated based on thechannel estimation. In step 915, an estimated frequency error signal isgenerated based on at least one of the finger weights. In step 920, arotating phasor signal is generated based on the estimated frequencyerror signal. In step 925, the rotating phasor signal is multiplied withthe sample data stream to correct the frequency of the sample datastream.

FIG. 10 is a flow diagram of a process 1000 including method steps forgenerating the estimated frequency error signal based on a finger weightextracted from Rake receiver finger weights generated by the channelestimator used in the process 900 of FIG. 9. In step 1005, anappropriate finger weight is extracted from the Rake receiver fingerweights (generated in step 910 of FIG. 9). In step 1010, the extractedfinger weight is delayed. In step 1015, a conjugate of the extractedfinger weight is generated. In step 1020, the conjugate of the extractedfinger weight is multiplied with the delayed extracted finger weight togenerate a phase difference signal which represents the phase differencebetween the conjugate extracted finger weight and the delayed extractedfinger weight. In step 1025, the phase difference between the conjugateextracted finger weight and the delayed extracted finger weight ismeasured. In step 1030, the estimated frequency error signal isgenerated by averaging the measured phase difference. In step 1035, theestimated frequency error signal is selectively prevented from beingupdated based on a value of a threshold signal.

Although the features and elements of the present invention aredescribed in the preferred embodiments in particular combinations, eachfeature or element can be used alone without the other features andelements of the preferred embodiments or in various combinations with orwithout other features and elements of the present invention.

1. A baseband frequency correction (BFC) apparatus comprising: anequalizer for performing equalization on a sample data stream andgenerating filter tap values based on the equalization; and a frequencyerror estimator for generating an estimated frequency error signal basedon at least one of the filter tap values, the frequency error estimatorcomprising: a tap extraction unit for extracting an appropriate tapvalue from the filter tap values generated by the equalizer andoutputting the extracted tap value; a delay unit coupled to the tapextraction unit for delaying the extracted tap value; a conjugategenerator coupled to the tap extraction unit and the delay unit forgenerating a conjugate of the extracted tap value; a first multipliercoupled to the delay unit and the conjugate generator for multiplyingthe conjugate of the extracted tap value with the delayed extracted tapvalue to generate a phase difference signal which represents the phasedifference between the conjugate extracted tap value and the delayedextracted tap value; an arctangent unit coupled to the output of thefirst multiplier for measuring the phase difference between theconjugate extracted tap value and the delayed extracted tap value; and afirst filter for averaging the measured phase difference, whereby theaveraged phase difference is output from the frequency error estimatoras the estimated frequency error signal.
 2. The apparatus of claim 1further comprising: an oscillator for generating a rotating phasorsignal based on the estimated frequency error signal.
 3. The apparatusof claim 2 further comprising: a multiplier for multiplying the rotatingphasor signal with the sample data stream to correct the frequency ofthe sample data stream.
 4. The apparatus of claim 3 wherein thefrequency error estimator further comprises a comparator circuit coupledto the first multiplier and the first filter for selectively sending ahold signal to the first filter based on a value of a threshold signal,wherein the hold signal causes the first filter to pause.
 5. Theapparatus of claim 4 wherein the comparator circuit comprises: amagnitude detector coupled to the first multiplier and the arctangentunit for calculating a magnitude of the phase difference signal; and acomparator for comparing the magnitude of the phase difference signalwith the value of the threshold signal and generating the hold signal ona condition that the magnitude of the phase difference signal is belowthe value of the threshold signal.
 6. The apparatus of claim 5 whereinthe comparator circuit further comprises: a second filter for averagingthe magnitude of the phase difference signal; and a second multiplierfor multiplying a scaling factor with the averaged magnitude of thephase difference signal to generate the threshold signal.
 7. Theapparatus of claim 4 wherein the comparator circuit comprises: a powerdetector for calculating instantaneous power of the phase differencesignal; and a comparator for comparing the instantaneous power of thephase difference signal with a value of a threshold signal, whereby thecomparator generates the hold signal on a condition that theinstantaneous power of the phase difference signal is below the value ofthe threshold signal.
 8. The apparatus of claim 7 wherein the comparatorcircuit further comprises: a second filter for averaging theinstantaneous power of the phase difference signal; and a secondmultiplier for multiplying a scaling factor with the averagedinstantaneous power of the phase difference signal to generate thethreshold signal.